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Author: admin, 14.11.2015. Category: Cause Of Ringing In Ears

Squarewave testing is a way to test many things at once, but you have to know what to look for. By using squarewaves to test equipment it's possible to see a wide range of potential problems, but you need to be able to determine what really is a problem and what is normal behaviour. If you've not done any squarewave testing, I encourage you to hook up a few circuits (they are shown in the appendix) so you can experiment. Before we start looking at modifications to the waveform, it's useful to see the squarewave and its harmonics. The spectrum shows that there is a continuous band of frequencies that are exact odd harmonics of the fundamental.
As the allowable output voltage swing increases, so too does the slew rate for a given frequency. When the input waveform is significantly faster than the amplifier stage, the leading and trailing edges will no longer be vertical, because the amplifying circuit has a limited bandwidth.
The waveform shows the same squarewave seen in Figure 1A, but with a 1k + 10nF capacitor arranged as a low pass filter. It's also worth examining the risetime - it's usually measured between 10% and 90% of the waveform's peak-to-peak amplitude. Figure 2 shows gentle bandwidth limiting, but if there is significant high frequency attenuation the squarewave starts to lose most of its 'character'. Simply by looking at the wave shape with a squarewave input, it is quite easy to see high frequency attenuation.
If you are testing tone controls, you fully expect to be able to get treble boost and cut, and the response should smoothly transition between them as the pot is rotated. The input for this was the same waveform shown in Figure 2, with a band limiting filter of 1k and 10nF at the input. If the top and bottom (the 'flat') parts of the squarewave are tilted, it means there is phase shift. Conversely, if the bass is boosted the flat section of the squarewave will rise, as shown below. This is one of the great benefits of squarewave tests - you can see variations both above and below the fundamental frequency, something that isn't possible with sinewave testing. The shape of the waveform is changed radically, and it no longer looks anything like a squarewave. With the values shown, the circuit will oscillate at any frequency from about 114Hz up to 1.25kHz.
The remaining five inverters are wired in parallel to provide the maximum possible output current and allows a fairly low output impedance. The minimum suggested supply voltage is +5V, or a 9V battery is a convenient way to power the circuit. Squarewave testing lets you see things that can be quite tedious to measure using 'traditional' sinewave tests. Using a squarewave is useless for checking crossover distortion or an amplifier's overload recovery time. There is plenty of info on the Net about squarewave testing, but sadly some people have no idea how to interpret the results.
The diode rectifies the ringing signal to supply the operating power to the audio relaxation oscillator made up of LI, L2, Rl, R2, and C.
How to stop nuisance calls PC AdvisorPC Advisor The main issue with nuisance calls is that even if you know not to answer them, it's still a pain that you have to listen to the at all, or get up and walk to it to find out who's calling. The circuit in Figure 1 performs active voltage-to-current conversion or acts as a variable-gain current mirror with high precision and bandwidth. A source of high frequency electricity is used to drive a large alternating current through a coil.
The passage of current through this coil generates a very intense and rapidly changing magnetic field in the space within the work coil.
In addition to this, the high frequency used in induction heating applications gives rise to a phenomenon called skin effect.
For ferrous metals like iron and some types of steel, there is an additional heating mechanism that takes place at the same time as the eddy currents mentioned above.
It is interesting to note that steel looses its magnetic properties when heated above approximately 700°C. Induction heating can be used for any application where we want to heat an electrically conductive material in a clean, efficient and controlled manner. One of the most common applications is for sealing the anti-tamper seals that are stuck to the top of medicine and drinks bottles. Another common application is "getter firing" to remove contamination from evacuated tubes such as TV picture tubes, vacuum tubes, and various gas discharge lamps. Yet another common application for induction heating is a process called Zone purification used in the semiconductor manufacturing industry.
The work coil is made to resonate at the intended operating frequency by means of a capacitor placed in series with it. This arrangement is commonly used in things like rice cookers where the power level is low, and the inverter is located next to the object to be heated. The tank capacitor is typically rated for a high voltage because of the resonant voltage rise experienced in the series tuned resonant circuit.
The work coil is made to resonate at the intended operating frequency by means of a capacitor placed in parallel with it. Work coils using this technique often consist of only a few turns of a thick copper conductor but with large currents of many hundreds or thousands of amps flowing. In the parallel resonant tank circuit the work coil can be thought of as an inductive load with a "power factor correction" capacitor connected across it. We can think of the tank circuit incorporating the work coil (Lw) and its capacitor (Cw) as a parallel resonant circuit. This has a resistance (R) due to the lossy workpiece coupled into the work coil due to the magnetic coupling between the two conductors.
In practice the resistance of the work coil, the resistance of the tank capacitor, and the reflected resistance of the workpiece all introduce a loss into the tank circuit and damp the resonance.
When driven at resonance the current drawn by the tank capacitor and the work coil are equal in magnitude and opposite in phase and therefore cancel each other out as far as the source of power is concerned. The job of the matching network is simply to transform this relatively large loss resistance across the tank circuit down to a lower value that better suits the inverter attempting to drive it. In the case of an L-match network it can transform the relatively high load resistance of the tank circuit down to something around 10 ohms which better suits the inverter. The squarewave voltage generated by most half-bridge and full-bridge circuits is rich in high frequency harmonics as well as the wanted fundamental frequency. Finally, with correct tuning the L-match network is able to provide a slight inductive load to the inverter. In summary, the inclusion of an L-match network between the inverter and the parallel resonant tank circuit achieves two things. Presentation of a rising inductive reactance to high frequency harmonics to keep the inverter safe and happy. Looking at the previous schematic above we can see that the capacitor in the matching network (Cm) and the tank capacitor (Cw) are both in parallel. This arrangement incorporates the work coil into a parallel resonant circuit and uses the L-match network between the tank circuit and the inverter.
Only comparatively low current flows along the transmission line from the inverter to the tank circuit, so this can use lighter duty cable. Any stray inductance of the transmission line simply becomes part of the matching network inductance (Lm.) Therefore the heat station can be located away from the inverter.
The inverter sees a sinusoidal load current so it can benefit from ZCS or ZVS to reduce its switching losses and therefore run cooler. The series matching inductor can be altered to cater for different loads placed inside the work coil. The tank circuit can be fed via several matching inductors from many inverters to reach power levels above those achievable with a single inverter. Another advantage of the LCLR work coil arrangement is that it does not require a high-frequency transformer to provide the impedance matching function.
The system schematic belows shows the simplest inverter driving its LCLR work coil arrangement. Note that this schematic DOES NOT SHOW the MOSFET gate-drive circuitry and control electronics! The inverter in this demonstration prototype was a simple half-bridge consisting of two MTW14N50 MOSFETs made my On-semiconductor (formerly Motorola.) It is fed from a smoothed DC supply with decoupling capacitor across the rails to support the AC current demands of the inverter. The DC-blocking capacitor is used merely to stop the DC output from the half-bridge inverter from causing current flow through the work coil. In high power designs it is common to use a full-bridge (H-bridge) of 4 or more switching devices. At still higher powers it is possible to use several seperate inverters effectively connected in parallel to meet the high load-current demands. These matching inductors also provide a number of additional benefits when inverters are paralleled in this way. The LCLR work coil arrangement is very well behaved under a variety of possible fault conditions. All of these failures result in an increase in the impedance being presented to the inverter and therefore a corresponding drop in the current drawn from the inverter.
The worst thing that can happen is that the tank circuit becomes detuned such that its natural resonant frequency is just above the operating frequency of the inverter.
Whilst not a problem for special fast recovery rectifiers, this forced recovery can cause problems if the MOSFETs intrinsic body diodes are used to provide the free-wheel diode function.
The power processed by the inverter can be decreased by reducing the supply voltage to the inverter.
It should be noted however, that the exact power throughput in kilowatts depends not only on the DC supply voltage to the inverter, but also on the load impedence that the work coil presents to the inverter through the matching network.
The power processed by the inverter can be decreased by reducing the on-time of the switches in the inverter. The power supplied by the inverter to the work coil can be reduced by detuning the inverter from the natural resonant frequency of the tank circuit incorporating the work coil.
In order to reduce the power throughput the inverter is normally detuned on the high side of the tank circuits natural resonant frequency. This method of controlling power level by detuning is very simple since most induction heaters already have control over the operating frequency of the inverter in order to cater for different workpieces and work coils.
The power supplied by the inverter to the work coil can be varied by altering the value of the matching network components. The matching network is responsible for transforming the load impedance of the workcoil to a suitable load impedance to be driven by the inverter. The power supplied by the inverter to the work coil can be varied in coarse steps by using a tapped RF power transformer to perform impedance conversion.
Additionally if the ferrite power transformer is placed between the inverter's output and the input to the L-match circuit its design constraints are relaxed in many ways. Finally the transformer design should be optimised for minimum inter-winding capacitance and good insulation at the expense of increased leakage inductance. When the work coil is driven by a voltage-fed full-bridge (H-bridge) inverter there is yet another method of achieving power control.
This technique is highly effective as power control can be achieved at the lower power control side. The requirements for capacitors used in high power induction heating are perhaps the most demanding of any type of capacitor. Correct choice of dielectrics and extended foil construction techniques are used to minimise the amount of heat generated and keep effective-series-inductance to a minimum. Note the large surface area of the connection plates on the Celem conduction-cooled components and the reactive power rating (KVAR) printed on the rating label. The LCLR network is a 3rd order resonant system consisting of two inductors, one capacitor and one resistor. From the amplitude part of the bode plot it can be seen that maximum voltage is developed across the work coil (top red trace) at one frequency only. From the bottom graph we can see that for frequencies below the maximum power point, the work coil voltage (green) is in-phase with the output voltage from the inverter. From the bottom graph we can also see that the load current from the inverter exhibits not one but two abrupt phase changes as the operating frequency is progressively increased. The bode plot above shows the area of interest around the null frequency and the maximum power point in more detail. From these graphs we can deduce a few implications for any control system that must track the resonant frequency of the LCLR arrangement and control power throughput. Conversely we can say that with a large lossy workpiece present, there will be less resonant rise inherent in the LCLR arrangement and the inverter will have to supply more load current in order to achieve the required level of current in the work coil. Finally a number of points are worthy of consideration from the plot above when considering an automatic control stratergy to track the resonant frequency of an LCLR induction heater. In contrast we can see that the work-coil voltage (and tank capacitor voltage) phase (bottom red plot) is monotonic with increasing frequency.
The white vertical line indicates the frequency at which the tank capacitor voltage (and also the work coil voltage) lag the inverter output voltage by 90 degrees. Looking to the right of the white line we have the area shaded in blue labelled "Inductive Load region." As the operating frequency is increased above the maximum power point, the voltage across the work coil decreases and less heating effect is generated in the workpiece. Conversely, to the left of the white line we have a band of frequencies labelled "Capacitive Load region." As the operating frequency is decreased below the maximum power point, the work coil voltage also falls and less heating effect takes place. The vertical purple line marks the other end of the capacitive load region, where the inverter load current transitions again to lagging "Inductive" load current. Note: The phase of the tank capacitor voltage was suggested as a control variable and discussed extensively in the plots above. This shows the inverter output current waveform when driving the LCLR work coil arrangement close to its resonant frequency. This shows the inverter output current waveform when driving the LCLR work coil arrangement substantially above its natural resonant frequency. This shows the voltage across the work coil under normal operation when driven close to resonance. This shows the output voltage from the inverter when it is mistuned to a frequency that is below the natural resonant frequency of the work coil. This shows the output voltage from the inverter when it is tuned very slightly above the natural resonant frequency of the work coil. This shows the output voltage from the inverter when it is tuned precisely to the resonant frequency of the work coil. This article explains the many different waveforms you can get from an amplifier or filter, and shows what each waveform actually means.
This is an excellent way to learn just how a squarewave can be modified by filters, including tone control circuits which are very enlightening.
In many cases you will see squarewave response that's far greater than shown in any of the diagrams.
This was deliberately simulated, using a squarewave with lightning-fast rise and fall times (1ns). By definition, the mark-space ratio is 1:1, meaning that positive half cycles and negative half cycles are exactly equal in amplitude and duration. I've only showed harmonics up to 160kHz - not because the simulator can't show more, but because there's no point. You don't need to know this, but it is useful background for you to use if you ever happen to need it. It is very easy to perform a squarewave test and end up with an entirely wrong answer if you're not careful. This dramatically changes the required slew rate, but it's very common (and advisable) to ensure that an amplifier can reproduce no less than 50% output voltage at 20kHz to ensure an acceptable safety margin. Before using the squarewave, it should be passed through a filter that rolls off the response above 20kHz. The reason for this is simple, in that many circuits will have some small 'disturbance' as the voltage starts to change and just before it reaches the opposite peak voltage. Feel free to use a higher order filter tuned to a higher frequency if it makes you feel any better.
Although the amplitude of the fundamental is not greatly affected, all the harmonics are reduced by the filter, producing the very rounded waveform shown.
The signal is then processed through a simple opamp circuit that applies 6dB of boost at the same frequency.
It always shows up in the response, but it can be missed as being just a bit of high frequency boost. The duration of the ringing gives you an idea of the resonant circuit's Q, and high-Q resonance will cause the oscillation to continue for several (or many) cycles.
The slope of the (normally) flat peaks show the effects of bass boost and cut, and the amount of slope is an instant indicator that bass frequencies are either attenuated or boosted. However, in almost all cases, phase shift comes free with response variations, and all cases of reduced or boosted bass response are accompanied by phase shift, so the tilt can be seen as an indicator of bass boost or cut. However, it's not the subtle variations of harmonic voltages that cause the waveform to change, but rather their relative phase that influences the shape of the waveform. If you test with a sinewave and want to see the effect at (say) 20Hz, you have to use a 20Hz sinewave.
However, the harmonic amplitudes are unchanged, and only their relative phase has been altered. If the phase shift network's frequency is increased, the trailing edge of the squarewave is affected as shown. If you look closely and compare it with Figure 9, you'll see that the difference is vanishingly small - that's because the crossover network introduces the same phase shift as the phase shift network.
This is a 'composite' of the various effects described, plus a representative indication of band pass and band stop filters (both low Q).


Note that the output current from CMOS devices is limited, so the circuit can't drive a low impedance load without the level being reduced dramatically. In particular, the frequency response can be seen at a glance - a squarewave with a frequency of between 400Hz and 1kHz will show any high frequency rolloff, ringing and low frequency rolloff.
This article, including but not limited to all text and diagrams, is the intellectual property of Rod Elliott, and is Copyright © 2015. Moreover, L2 together with Ql acts as an opto-isolator, totally isolating the telephone line from the rest of the circuit.
This results in frequency modulation of the audio oscillator giving it an attention-getting warble.
A typical application is testing high-speed ICs or other devices that have inputs designed to be driven from current-steering DACs to enable a modulated voltage source to control the devices.
The arrangement of the work coil and the workpiece can be thought of as an electrical transformer. The intense alternating magnetic field inside the work coil repeatedly magnetises and de-magnetises the iron crystals. A foil seal coated with "hot-melt glue" is inserted into the plastic cap and screwed onto the top of each bottle during manufacture. A ring of conductive material called a "getter" is placed inside the evacuated glass vessel.
This is a process in which silicon is purified by means of a moving zone of molten material. For example, an impedance matching network is often required between the High Frequency source and the work coil in order to ensure good power transfer.
The main drawbacks of the series resonant arrangement are that the inverter must carry the same current that flows in the work coil.
It must also carry the full current carried by the work coil, although this is typically not a problem in low power applications. The PFC capacitor provides reactive current flow equal and opposite to the large inductive current drawn by the work coil. This refers to the electronics that sits between the source of high frequency power and the work coil we are using for heating.
Therefore it is useful to combine all of these losses into a single "loss resistance." In the case of a parallel resonant circuit this loss resistance appears directly across the tank circuit in our model.
This means that the only load seen by the power source at the resonant frequency is the loss resistance across the tank circuit. This figure is typical to allow the inverter to run from several hundred volts whilst keeping currents down to a medium level so that standard switch-mode MOSFETs can be used to perform the switching operation. The inductor at the input to the L-match network presents a progressively rising inductive reactance to all frequencies higher than the resonant frequency of the tank circuit.
Direct connection of such a voltage source to a parallel resonant circuit would cause excessive currents to flow at all harmonics of the drive frequency! The matching network is used to make the tank circuit appear as a more suitable load to the inverter, and its derivation is discussed in the section above.
The large circulating current is confined to the work coil and its parallel capacitor, which are usually located very close to each other. The matching inductors provide inherent sharing of the load current between the inverters and also make the system tolerant to some mismatching in the switching instants of the paralleled inverters.
Ferrite transformers capable of handling several kilowatts are large, heavy and quite expensive. However, it should be realised that the quality and regulation of the power supply for induction heating applications is not critical.
It is sized sufficiently large that it does not take part in the impedance matching, and does not adversely effect the operation of the LCLR work coil arrangement.
In such designs the matching inductance is usually split equally between the two bridge legs so that the drive voltage waveforms are balanced with respect to ground. However, the seperate inverters are not directly tied in parallel at the output terminals of their H-bridges.
Firstly, the impedance BETWEEN any two inverter outputs is equal to twice the value of the matching inductance. These large current spikes still represent a significant power loss and threat to reliability. This can be done by running the inverter from a variable voltage DC supply such as a controlled rectifier using thyristors to vary the DC supply voltage derived from the mains supply.
Therefore if precise power control is required the actual induction heating power must be measured, compared to the requested "power setting" from the operator and an error signal fed back to continually adjust the DC link voltage in a closed-loop fashion to minimise the error. As the operating frequency of the inverter is moved away from the resonant frquency of the tank circuit, there is less resonant rise in the tank circuit, and the current in the work coil diminishes.
This causes the inductive reactance at the input of the matching circuit to become increasingly dominant as the frequency increases.
The L-match network between the inverter and the tank circuit technically consists of an inductive and a capacitive part. Altering the inductance of the matching inductor adjusts the value to which the load impedance is translated. There is a also a shift in the resonant frequency of the overall system - This is the price to pay for combining the L-match capacitance and tank capacitance into one unit.
Although most of the benefit of the LCLR arrangement is in the elimination of a bulky and expensive ferrite power transformer, it can cater for large changes in system parameters in a way that is not frequency dependent.
Firstly, locating the transformer in this position means that the impedances at both windings are relatively high. The reason for this is that any leakage inductance exhibited by a transformer located in this position merely adds to the matching inductance at the input to the L-match circuit. If the switching instants of both bridge legs can be controlled independently then it opens up the possibility of controlling power throughput by adjusting the phase shift between the two bridge legs.
This means there is no voltage across the work coil arrangement and no current flows through the work coil. The power factor seen by the inverter always remains good because the inverter is not detuned from the resonant frequency of the work coil, therefore reactive current flow through free-wheeling diodes is minimised.
The capacitor bank used in the tank circuit of an induction heater must carry the full current that flows in the work coil for extended periods of time. The high operating frequency causes significant losses due to dielectric heating and due to skin effect in the conductors.
However, even with these techniques Induction heating capacitors still exhibit significant power dissipation due to the enormous RF currents they must carry.
Higher power units pictured above in aluminium cases have connections for water cooling hoses to remove the heat generated internally. The bode plot below shows the way in which some of the voltages and currents within the network change as the drive frequency is altered. At this frequency current through the work coil is also maximum and the largest heating effect is developed at this frequency.
As the operating frequency increases the phase angle of the work coil voltage changes abruptly through 180 degrees (phase inversion) right at the point where maximum power is being processed. Inverter load current initially lags the inverter's output voltage by 90 degrees at low frequencies. It also shows a family of curves depicting the behaviour of the induction heating tank circuit with a variety of different workpieces present. This gives rise to the sharply peaking currents and voltages in the top graph, and the abruptly changing phase shifts in the bottom graph.
Firstly there is more resonant rise in the LCLR network when there is no workpiece present. However, the control electronics now do not need to track the resonant frequency so closely since the diminished Q gives a load current that shifts phase in a more leisurely manner. For very lossy workpiece materials, (or large volumes of metal that introduce a significant overall loss) we can see that the inverter load current phase (bottom green plot) sometimes fails to ever cross through zero degrees to leading phase.
Furthermore it consistently passes through the -90 degree phase-lag point exactly at the frequency which gives maximum power regardless of how heavily the work coil is loaded. This is also the point where maximum voltage is developed across the work coil and maximum current flows through it. The inverter load current also falls and begins to lag in phase relative to the output voltage of the inverter. However, this is accompanied by the inverter load current possibly slewing to a leading phase angle when losses in the workpiece are low and Q factor is high. This second Inductive region is of little interest since it does not achieve significant power throughput, and cannot be reached without passing through the potentially damaging capacitive load region anyway. This is because this voltage can be easily sensed using a high-frequency voltage transformer and provides all the necessary control information.
Notice the very fast rise and fall times of the squarewave accompanied by excessive voltage overshoot and ringing. Notice that the rise and fall times of the squarewave are more controlled, and there is comparatively little overshoot or ringing. Although this situation actually achieves maximum power throughput, it does not quite achieve Zero Voltage Switching of the MOSFETs. A squarewave is a signal that's rich in harmonics, and because it's symmetrical the harmonics are all odd - 1st (the fundamental), 3rd, 5th, etc.
That doesn't indicate a fault, it simply means that the circuit has more of whatever is modifying the waveform or phase than these examples show. The RMS value of a squarewave is simply its peak value, so for the waveform shown it's exactly 1V RMS. For the most part you'll probably never need it, so I wouldn't worry about trying to remember the formula. TIM may have been discredited (along with its siblings), but it doesn't make any sense to limit an amplifier if it's not necessary.
Failure to use bandwidth limiting won't hurt the amplifier, but you may see artefacts that will not appear in normal use. This will be the case for all subsequent waveforms, and where possible the same vertical scale will be used as well. By excluding the fist and last 10% of the waveform these disturbances are minimised and the true risetime (and from that the slew rate) can be determined more accurately.
A squarewave also lets us see if there is high frequency boost, as we see in the next graph. You might expect the two to cancel out (which can also be done), but because the cut and boost circuits are not perfectly symmetrical we see a small amount of treble boost. By testing with a squarewave you can instantly see things that are otherwise easily missed.
Some very steep filters can also cause damped ringing as well, especially Chebychev types which have a peak before they roll off. It's important to verify that the slope on both the positive and negative parts of the waveform are perfectly complementary.
More bass boost causes the angle of slope to increase, and it doesn't take long for anyone with a squarewave generator, oscilloscope and a set of bass and treble controls to see the interactions that are possible. With a squarewave, you can see the effect of boost or cut at frequencies well below the test frequency.
You would expect this waveform to sound completely different from the original squarewave, but it doesn't.
A frequency scan will not show any anomalies with anything that affects phase but leaves the frequency response unchanged. These simulations both used filters providing 6dB boost or cut at the resonant frequency, and the squarewave input is at the same frequency as resonance.
You need only to supply a sinewave at the required frequency and the CMOS IC will convert it to a very good squarewave with extremely fast risetime. This alternative input stage means that you don't need a sinewave oscillator, as you can generate squarewaves directly. This is still a fairly severe test, but it's better than using a signal that is dramatically faster than any audio signal.
The measurement doesn't give you actual numbers, but it is very quick and instantly alerts you if anything is out of the ordinary. Likewise, you can't test distortion with a squarewave, because the waveform itself has a (theoretical) total harmonic distortion (THD) of over 48%.
It's especially useful with graphic equalisers because you may have a great many sliders, and you can test each in turn and listen to the result. Reproduction or re-publication by any means whatsoever, whether electronic, mechanical or electro- mechanical, is strictly prohibited under International Copyright laws. The oscillator audio frequency is optically coupled to the photo-Darlington which drives Q2 and thus the speaker. Since it is non-contact, the heating process does not contaminate the material being heated. The work coil is like the primary where electrical energy is fed in, and the workpiece is like a single turn secondary that is short-circuited. This rapid flipping of the magnetic domains causes considerable friction and heating inside the material. This means that above 700°C there can be no heating of the material due to hysteresis losses. These foil seals are then rapidly heated as the bottles pass under an induction heater on the production line. Since induction heating is a non-contact process it can be used to heat the getter that is already sealed inside a vessel. An Internet Search is sure to turn up more details on this process that I know little about.
Water cooling systems are also common in high power induction heaters to remove waste heat from the work coil, its matching network and the power electronics. The series resonance also magnifies the voltage across the work coil, far higher than the output voltage of the inverter alone. In addition to this the voltage rise due to series resonance can become very pronounced if there is not a significantly sized workpiece present in the work coil to damp the circuit.
The parallel resonance also magnifies the current through the work coil, far higher than the output current capability of the inverter alone.
The key thing to remember is that this huge current is localised to the work coil and its capacitor, and merely represents reactive power sloshing back-and-forth between the two. In order to heat a solid piece of metal via induction heating we need to cause a TREMENDOUS current to flow in the surface of the metal. This resistance represents the only component that can consume real power, and therefore we can think of this loss resistance as the load that we are trying to drive power into in an efficient manner.
This is very important when the work coil is to be fed from a voltage-source inverter that generates a squarewave voltage output.
This is because the tank capacitor in the parallel resonant circuit would present a progressively lower capacitive reactance to increasing frequencies. In addition to this, the transformer must be cooled to remove excess heat generated by the high currents flowing in its conductors.
The DC-blocking capacitor can also be eliminated if current mode control is used to ensure that no net DC flows between the bridge legs. Each of the distributed inverters is connected to the remote work coil via its own pair of matching inductors which ensure that the total load is spread evenly among all of the inverters. This inductive impedance limits the "shoot between" current that flows between paralleled inverters if their switching instants are not perfectly synchronised. However, it should be realised that proper control of the inverter operating frequency should ensure that it tracks the resonant frequency of the tank circuit. The impedance presented to the inverter is largely constant with varying power level, so the power throughput of the inverter is roughly proportional to the square of the supply voltage. This is necessary to maintain constant power because the resistance of the workpiece changes considerably as it heats up. The load current is then left to freewheel through the devices body diodes during the deadtime when both devices are turned off. Therefore less circulating current is induced into the workpiece and the heating effect is reduced. Therefore the current drawn from the inverter by the matching network starts to lag in phase and diminish in amplitude. But the capacitive part is in parallel with the work coil's own tank capacitor, and in practice these are usually one and the same part. In general, decreasing the inductance of the matching inductor causes the work coil impedance to be transformed down to a lower impedance.
The L-match network essentially borrows some of the capacitance from the tank capacitor to perform the matching operation, thus leaving the tank circuit to resonate at a higher frequency. The ferrite power transformer can also provide electrical isolation as well as performing impedance transformation duty to set the power throughput.
Conversely, when both bridge legs switch in anti-phase maximum current flows through the work coil and maximum heating is achieved.
Finally stray inductance must be kept to an absolute minimum so that the capacitor appears as a lumped circuit element compared to the reasonably low inductance of the work coil it is connected to.
Therefore an important factor in their design is allowing the effective removal of heat from within the capacitor to extend the life of the dielectric. The GREEN traces represent the current passing through the matching inductor, and therefore the load current seen by the inverter.
It can be seen that this frequency corresponds to the maximum load current drawn from the inverter (top green trace.) It is worth noting that the magnitude of the inverter load current has a null at a frequency only slightly lower than that which gives maximum heating. The phase angle of the work coil voltage then remains shifted by 180 degrees from the inverter output voltage for all frequencies above the maximum power point.
Load current abruptly slews through 180 degrees to a phase lead of 90 degrees as the operating frequency passes through the "null frequency" of the network. This allows us to get a feel for how the network behaves with a large lossy workpiece to having no workpiece present at all, and all loads in between.


Therefore the current delivered from the inverter should be decreased to prevent the work coil and tank capacitor currents sky-rocketing in the absence of any significant loss in the system.
This means that the inverter load current with heavy workloads cannot be in-phase and is always lagging by some amount.
The white line is where you want to be to develop the maximum possible heating effect in the workpiece. These properties make the blue shaded region the ideal place to operate in order to achieve control over induction heating power. This is undesirable for many solid-state inverters as the leading load current causes loss of ZVS and leads to forced reverse-recovery of free-wheeling diodes incurring raised switching losses and voltage overshoots.
When the LCLR network is driven from a squarewave inverter voltage there is also risk of significant current flow at a harmonic of the drive frequency. Whilst it exhibits a 90 degree phase shift relative to the inverter output voltage (which may at first appear undesirable) it is still a better control variable than trying to sense the tank capacitor current. At frequencies above the natural resonant frequency of the LCLR work coil arrangement the inductive reactance of the matching network dominates and the inverter's load current lags the applied voltage.
This is also true for the current waveform and minimises harmonic radiation and RF interference.
These are all attributed to forced reverse-recovery of the MOSFET body diodes whilst enduring this undesirable operating mode. This is due to the Zero Voltage Switching (ZVS) which takes place when the inverter runs in this favourable operating mode. A 'perfect' squarewave has harmonics that extend to infinity, but somewhat predictably perfection isn't attainable. For example, it's easy to get far more treble cut or boost than you'll see in any of the following examples, so you have to learn how to interpret the results. A rise and fall time of 1ns indicates a bandwidth well in excess of 1GHz, and this is not necessary or desirable for amplifier testing.
When testing any audio device, you must be aware of the simple fact that music does not contain very fast risetime signals, and most media (vinyl, CD, etc.) are actually not very demanding. It also doesn't make sense to go to a great deal of additional effort to design an amplifier that can reproduce full power at 100kHz (or even 20kHz), because it will never be needed. A low-pass filter using a 1k resistor and 10nF capacitor gives a response that's considerably faster than the harmonic structure of music, but doesn't stress any amplifier too hard. From this waveform, you can see the result of a low pass filter - the risetime is increased.
This will decrease the risetime only slightly (about 19us), but the sharp cutoff slope will cause slight overshoot of the leading edge. This is commonly referred to as 'overshoot', because the leading edge of the waveform extends beyond the steady state value. Remember that these waveforms all use the band limited squarewave shown in Figure 2, and a faster risetime will make the effects more apparent. Valve (vacuum tube) output transformers are another source of ringing, caused by the natural resonant frequency of the transformer. If the waveform shows a rising slope (as seen in Figure 8) that always means there is some bass boost. If the bass cut is made much more severe, the slope is exaggerated as shown when a bass cut filter has a frequency that's higher then the squarewave frequency.
Certainly the harmonics change, but it's possible to change the wave shape below the fundamental frequency, which seems as though it should not be possible.
In fact, it sounds just like the original in all respects, and you can even vary the amount of phase shift while listening, and hear no change at all - provided it's done slowly. Probably one of the most extreme is a crossover network, which ideally leaves the frequency response unchanged when the high and low pass outputs are summed, but the modified phase response is seen clearly and instantly. If you know what you are listening to, you will hear a difference whether there is a change or not. You will see different effects again as the test frequency is changed relative to the filter frequency.
Excellent supply bypassing is essential, otherwise the waveform will be corrupted by supply variations created when the CMOS inverters switch. The whole circuit is powered from a single 15V supply, but lower voltages can be used if more convenient. Needless to say, C2 can be omitted (or switched) so a very fast risetime is available if needed. Should you test both channels of a stereo amp and one is quite different from the other, you know immediately that something is wrong and requires further investigation. The harmonic amplitudes are affected by the equaliser, and you can both see and hear this easily. Unfortunately, there is just too much to even try to explain everything, but it's hoped that the info provided is enough to get the reader interested, and explains the results in enough detail so the results make sense. The author (Rod Elliott) grants the reader the right to use this information for personal use only, and further allows that one (1) copy may be made for reference. But after less than a year, Otto found out his old job in Indiana was open and he requested to be sent back this way.
Heating due to this mechanism is known as Hysteresis loss, and is greatest for materials that have a large area inside their B-H curve. An induction work coil is located close to the getter on the outside of the vacuum tube and the AC source is turned on.
This minimises losses in the inverter by allowing it to benefit from either zero-voltage-switching or zero-current-switching depending on the exact arrangement chosen. The inverter sees a sinusoidal load current but it must carry the full current that flows in the work coil.
Therefore the only real current flow from the inverter is the relatively small amount required to overcome losses in the "PFC" capacitor and the work coil. The incorporation of the L-match network into the LCLR work coil arrangement removes the necessity of a transformer to match the inverter to the work coil, saving cost and simplifying the design. Secondly, this same inductive reactance between inverters limits the rate at which fault current rises if one of the inverters exhibits a device failure, potentially eliminating failure of further devices. Therefore the only part of the matching network that is available to adjust is the inductor. This lower load impedance being presented to the inverter causes more power to be sourced from the inverter. For this reason the matching inductor is usually fixed or adjusted in coarse steps to suit the intended workpiece to be heated, rather than provide the user with a fully adjustable power setting.
The RED traces represent the voltage across the tank capacitor, which is the same as the voltage across the induction heating work coil.
Inverter current remains leading by 90 degrees until the maximum power point is reached, where it again abruptly slews through 180 degrees and returns to the 90 degree lagging phase once again. This causes less resonant rise in the inverter load current and the voltage across the work coil.
Secondly, the inverter load current with no load must be tracked very accurately if the inverter is not to see either a leading or lagging load current because it slews so quickly through zero degrees. In conclusion the inverter frequency should be controlled so as to achieve a consistent 90 degree lag between the tank capacitor voltage and the inverter output voltage in order to achieve maximum power throughput. If we look at the inverter load current phase (bottom green plot) we can see that this is always between 0 degrees and -90 degrees when it crosses the white line no matter how abruptly or slowly it slews.
By detuning the inverter drive frequency on the high-side of the maximum power point, power throughput can be reduced and the inverter always sees a lagging power factor. Therefore the capacitive load region is not recommended for achieving power throughput control. Although the tank capacitor current is in-phase with the inverter output this current can be many hundreds of amps making closed-core ferrite CTs impractical. Notice the triangular load current caused by the inductive load integrating the inverter's squarewave voltage output over time. In this case the voltage across the work coil is also higher than the DC bus voltage supplied to the inverter. These occur because the mid-point of the bridge leg has not been fully commutated to the opposite supply rail during the dead-time before the next MOSFET turns on. However, most common squarewave generators can easily produce harmonics that exceed the response capability of any audio amplifier. By themselves, they are decidedly uninteresting, but after processing through a filter or an amplifier we can see the essential characteristics of the device under test (DUT). It's important to understand that a 'true' squarewave is perfectly symmetrical so has no even order harmonics (2nd, 4th, etc.). So, just by looking at the squarewave response, you can see if there's high frequency rolloff, boost, or as we see next, ringing. This depends on the circuit of course - if it has its own bandwidth limitations what you see will be a combination of everything that affects the response.
Provided the ringing frequency is well above the audio band it usually doesn't cause any problems, but it should always be checked to ensure that the effects aren't audible. You are more likely to see asymmetrical waveforms with valve amps than anything using transistors or opamps. It's theoretically possible for phase shift alone to cause this, but I've never seen it in a working circuit. The range of waveforms you can see on the oscilloscope is vast, and it's simply impossible to try to show every case. Output rise and fall times are quoted as 40ns (typical) and 80ns (maximum) when running at 15V. The 100nF cap (C5) should be multilayer ceramic, and placed as close to the IC's supply pins as possible. After power is applied, it will take over 30 seconds before the squarewave is properly centred around zero volts. The filtered squarewave used for the tests described here has a THD of 43.5% according to the simulator. Ultimately, interpreting the results takes experience so I encourage readers to experiment with as many different circuits as they can so that interpretation becomes easy.
This can be contrasted with other heating methods where heat is generated in a flame or heating element, which is then applied to the workpiece. This can be a large contributing factor to the heat generated during induction heating, but only takes place inside ferrous materials. This makes heating steel above 700°C more of a challenge for the induction heating systems.
When the cap is removed, the foil remains providing an airtight seal and preventing any tampering or contamination of the bottle's contents until the customer pierces the foil. Within seconds of starting the induction heater, the getter is heated white hot, and chemicals in its coating react with any gasses in the vacuum.
The sinusoidal waveform at the work coil also represents a more pure signal and causes less Radio Frequency Interference to nearby equipment. For this reason the work coil often consists of many turns of wire with only a few amps or tens of amps flowing.
However, in this case it only has to carry the part of the load current that actually does real work. There is always some loss in this tank circuit due to dielectric loss in the capacitor and skin effect causing resistive losses in the capacitor and work coil. The inverter generally works better (and the design is somewhat easier) if it operates at fairly high voltage but a low current.
It results in large current spikes at the switching transitions as the inverter tries to rapidly charge and discharge the tank capacitor on rising and falling edges of the squarewave.
However, the designer should appreciate that a 1:1 isolating transformer may still be required between the inverter and the input to the LCLR work coil arrangement if electrical isolation is necessary from the mains supply. Finally, since all distributed inverters are already connected via inductors, any additional inductance between the inverters merely adds to this impedance and only has the effect of slightly degrading current sharing. However, a significant drawback of this method is the commutation of heavy currents between active devices and their free-wheel diodes.
In addition to this the lagging power factor ensures that the devices in the inverter still turn on with zero voltage across them, and there are no free-wheel diode recovery problems. Conversely, increasing the inductance of the matching inductor causes a higher load impedance to be presented to the inverter. The top graph shows the AC magnitudes of these two quantities, whilst the bottom graph shows the relative phase of the signals relative to the AC output voltage from the inverter. Therefore direct feedback from a Current Transformer (CT) on the inverter output is not a viable option.
This means that the inverter always sees a load current that is either in-phase or at worst slightly lagging in power factor.
Furthermore the 90 degree phase shift of the tank capacitor voltage waveform means that it's zero crossings are intentionally displaced in time away from the potentially noisy switching instants of the inverter.
Both of these properties are attributed to the high-Q factor of the induction heating tank circuit. In practice a small amount of inductive reactance presented to the inverter helps provide the required commutating current and achieve ZVS.
If your squarewave has any evidence of even order distortion, that means it's not symmetrical - the positive and negative portions of the waveform are unequal. This means that the actual level at 20kHz will typically be 20dB lower than the level at midrange frequencies. My function generator has a risetime of 12ns for a 1V RMS squarewave - much too fast for even the most esoteric amplifier, so a filter is needed to prevent the DUT from slew rate limiting.
Equally obvious is any circuit that applies high frequency boost, but we'll look at that in the next section. This effect is commonly seen at the output of Class-D amplifiers where the terminating impedance is not a perfect match for the filter components. This is difficult to understand based simply on a written explanation, and I'm not about to write pages of mathematical proof. Fortunately it's not necessary because the general trends will be obvious when you start to experiment for yourself. The input stage on the left will convert a sinewave into a squarewave, and the one on the right generates a squarewave directly. Squarewave testing is an additional tool, but it doesn't replace any of the more common tests that you might use.
For this reason ferrous materials lend themselves more easily to heating by induction than non-ferrous materials.
The result is that the getter absorbs any last remaining traces of gas inside the vacuum tube and increases the purity of the vacuum.
Therefore a small current is always drawn from the inverter even with no workpiece present. The inclusion of the L-match network between the inverter and the tank circuit negates this problem. This depends whether isolation is important, and whether the main PSU in the induction heater already provides sufficient electrical isolation to meet these safety requirements.
Therefore the distributed inverters for induction heating need not necessarily be located physically close to each other. Forced reverse recovery of the free-wheel diodes that can occur when the duty ratio is considerably reduced. Likewise the phase of the inverter current waveform and the work coil voltage slew less rapidly for lower Q factors. Whilst it may appear to work fine with no workpiece fitted or only moderate heating loads, it does not track the resonant frequency correctly and will fail to operate satisfactorily as the workload increases and network Q falls! Such a situation is ideal for supporting ZVS soft-switching in the inverter and preventing free-wheel diode reverse-recovery problems. This -90 degree phase shift of the voltage feedback signal can be allowed for in the design of the control electronics and is a small price to pay for the eased sensing and increased noise immunity gained.
For this reason the situation described for the previous photograph is preferable to being precisely in tune.
Some of these tests can be confronting, so an open mind (and a blind test regime) is essential. This is very significant since power factors in induction heating applications are typically low.
When a lossy workpiece is inserted into the work coil, this damps the parallel resonant circuit by introducing a further loss into the system.
Now the output of the inverter sees the inductive reactance of Lm in the matching network first, and all harmonics of the drive waveform see a gradually rising inductive impedance. If isolation transformers are included in the designs then they need not even run from the same supply! For this reason duty ratio control is not usually used in high power induction heating inverters. This property of the parallel resonant circuit can make a tenfold reduction in the current that must be supported by the inverter and the wires connecting it to the work coil. Therefore the current drawn by the parallel resonant tank circuit increases when a workpiece is entered into the coil. This means that maximum current flows at the intended frequency only and little harmonic current flows, making the inverter load current into a smooth waveform. If your hearing aid has a "T" switch, you can ask your company about getting a with an  17 May 2016 Check your ringer at the door Nevada Herald at the doorIn a recent discussion with some friends, we covered the topic of how "smart ," should be used in public schools. Our conversations covered issues, ranging from -- should they be allowed, to the many problems they pose.



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